Phase estimation for coherent optical detection

ABSTRACT

The present invention is a method and apparatus to make an estimate of the phase of a signal relative to the local oscillator in an optical coherent detection subsystem that employs a digital signal processor having a parallel architecture. The phase estimation method comprises operations that do not use feedback of recent results. The method includes a cycle count function so that the phase estimate leads to few cycle slips. The phase estimate of the present invention is approximately the same as the optimal phase estimate.

RELATED APPLICATIONS

This utility application claims the benefit of U.S. Provisional PatentApplication Ser. No. 60/676631 by Michael G. Taylor, filed Apr. 29,2005, and is hereby incorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to optical data transmission over optical fibers.Specifically, and not by way of limitation, the present inventionrelates a method and system providing an estimate of the phase of anoptical signal during coherent detection.

2. Description of the Related Art

A. Optical Fiber Communications

Information has been transmitted over optical fibers for some time.Details about this field are disclosed in “Optical CommunicationSystems,” by J. Gowar (Prentice Hall, 2nd ed., 1993) and “Fiber-opticcommunication systems” by G. P. Agrawal (Wiley, 2nd ed., 1997), whichare herein incorporated by reference. The information is usually in theform of binary digital signals, i.e. logical “1”s and “0”s, but fiberoptics is also used to transport analog signals, such as cable TVsignals. Every optical data transmission system has a transmitter, whichemits light modulated with information into the fiber, and a receiver atthe far end which detects the light and recovers the information. A longdistance digital link may also use one or more digital regenerators atintermediate locations. A digital regenerator receives a noisy versionof the optical signal, makes decisions as to what sequence of logicalvalues (“1”s and “0”s) was transmitted, and then transmits a cleannoise-free signal containing that information forward towards thedestination.

In the 1990's, optical amplifiers were deployed in telephony and cableTV networks, in particular erbium doped fiber amplifiers (EDFAs) weredeployed. These devices amplify the optical signals passing throughthem, and overcome the loss of the fiber without the need to detect andretransmit the signals. A typical long distance fiber optic digital linkmight contain some digital regenerators between the information sourceand destination, with several EDFAs in between each pair of digitalregenerators. Also in the 1990's, wavelength division multiplexing (WDM)was commercially deployed, which increased the information carryingcapacity of the fiber by transmitting several different wavelengths inparallel.

Digital communication systems may include forward error correction(FEC). This field is described in “Error Control Coding: From Theory toPractice” by Peter Sweeney (Wiley, 2002), which is herein incorporatedby reference. Block FEC codes are often used with fiber optictransmission systems. At the transmitter, a block of k symbols is codedto a longer block of n symbols, (i.e., overhead is added to the signal).The FEC code is called an (n,k) code. Then, at the receiver, the blockof n symbols is decoded back to the original k symbol block. Providedthe bit error rate (BER) is not too high, most of the bit errorsintroduced by the communications channel are corrected. Many fiber optictransmission systems use a standard FEC code based on the Reed-Solomon(255,239) code, as described in ITU-T Standard G.975 “Forward errorcorrection for submarine systems” (International TelecommunicationUnion, 2000).

B. Direct Detection & Coherent Detection

The transmitter unit for a single WDM channel contains a light source,usually a single longitudinal mode semiconductor laser. Information isimposed on the light by direct modulation of the laser current, or byexternal modulation, that is by applying a voltage to a modulatorcomponent that follows the laser. The receiver employs a photodetector,which converts light into an electric current. There are two ways ofdetecting the light: direct detection and coherent detection. All theinstalled transmission systems today use direct detection. Although itis more complex, coherent detection has some advantages, and it washeavily researched into in the 1980s and the start of the 1990s, and hasbecome of interest once again in the past few years.

Most deployed transmission systems impose information on the amplitude(or intensity, or power) of the signal. The light is switched on totransmit a “1” and off to transmit a “0”. In the case of directdetection, the photodetector is presented with the on-off modulatedlight, and consequently the current flowing through it is a replica ofthe optical power. After amplification the electrical signal is passedto a decision circuit, which compares it to a reference value. Thedecision circuit outputs an unambiguous “1” or “0”.

The coherent detection method treats the optical wave more like radio,inherently selecting one wavelength and responding to its amplitude andphase. “Fiber-optic communication systems” by G. P. Agrawal provides anintroduction to coherent detection. Coherent detection involves mixingthe incoming optical signal with light from a local oscillator (LO)laser source. FIG. 1 illustrates an example of a coherent receiversuitable for detecting a binary phase shift keyed (BPSK) signal. Theincoming signal 101 is combined with light 102 from a continuous wave(c.w.) local oscillator in a passive 2:1 combiner 103. The LO light hasclose to the same state of polarization (SOP) as the incoming signal andeither exactly the same wavelength (homodyne detection) or a nearbywavelength (heterodyne detection). When the combined signals aredetected at photodetector 104, the photocurrent contains a component ata frequency which is the difference between the signal and localoscillator optical frequencies. This difference frequency component,known as the intermediate frequency (IF), contains all the information(i.e., amplitude and phase) that was on the optical signal. Because thenew carrier frequency is much lower, typically a few gigahertz insteadof 200 THz, all information on the signal can be recovered usingstandard radio demodulation methods. Coherent receivers see only signalsclose in wavelength to the local oscillator, and so by tuning the LOwavelength, a coherent receiver can behave as though having a built-intunable filter. When homodyne detection is used, the photocurrent is areplica of the information and may be amplified 107 and then passed tothe decision circuit 106 which outputs unambiguous “1” or “0” values.With heterodyne detection, the photocurrent must be processed by ademodulator 105 to recover the information from the IF. FIG. 1illustrates a configuration for single-ended detection. There are otherconfigurations for coherent detection. For example, a balanced detectionconfiguration is obtained by replacing the 2:1 combiner by a 2:2combiner, each of whose outputs are detected and the difference taken bya subtracting component.

Following is a mathematical description of the coherent detectionprocess. (The complex notation for sinusoids is summarised in theAppendix.) The electric field of the signal may be written asRe└E_(s)(t)e^(iω) ^(s) ^(t+iφ) ^(s) ^((t))┘where E_(s)(t) is the slowly varying envelope containing the informationencoded on amplitude and phase of the optical signal, ω_(s) is theangular frequency of the optical carrier, and φ_(s)(t) is the slowlyvarying phase noise associated with the finite linewidth of the laser.Writing the phase noise separate from the modulation envelope E_(s)(t)has the advantage that in the case of digital information transmissionE_(s)(t) takes on only a small number of possible values, depending onthe digital signal format. Similarly, the electric field of the localoscillator is written asRe└E_(LO)e^(iω) ^(LO) ^(t+iφ) ^(LO) ^((t))┘where E_(LO) is a constant given that the local oscillator is c.w.,ω_(LO) is the angular frequency of the LO, and φ_(LO) (t) is the phasenoise on the LO. The electric fields of the signal and LO are written asscalar quantities because it is assumed that they have the same state ofpolarization. The electric field of the light arriving at thephotodetector 104 in FIG. 1 is the sum of the two electric fieldsE ₁ =Re└E _(s)(t)_(e) ^(i(ω) ^(s) ^(t+φ) ^(s) ^((t))) +E _(LO) e ^(i(ω)^(LO) ^(t+φ) ^(LO) ^((t)))┘and the optical power isP ₁ =E ₁ *E ₁P ₁ =|E _(s)(t)|² +|E _(LO)|²+2Re[E _(s)(t)E _(LO) *e ^(i(ω) ^(s) ^(−ω)^(LO) ^()t+i(φ) ^(s) ^((t)−φ) ^(LO) ^((t)))┘  (1)In the case of single ended detection only one output of the combiner isused. |E_(LO)|² is constant with time. |E_(s)(t)|² is small given thatthe local oscillator power is much larger than the signal power, and forphase shift keying (PSK) and frequency shift keying (FSK) modulationformats |E_(s)(t)|² is constant with time. The dominant term in equation1 is the beat term Re└E_(s)(t)E_(LO)*e^(i(ω) ^(s) ^(−ω) ^(LO) ^()t+i(φ)^(s) ^((t)−φ) ^(LO) ^((t)))┘. In appropriate conditions the beat termcan be readily obtained from the photocurrent in the single-endeddetection case. Alternatively, when |E_(s)(t)|² is not small and varieswith time, the beat term is produced directly by the balanced detectionconfiguration. The equations that follow refer to the beat term. It isassumed that this term is obtained by single ended detection given thatthe other terms do not contribute or by balanced detection.

There are two modes of coherent detection: homodyne and heterodyne. Inthe case of homodyne detection the frequency difference between signaland local oscillator is zero, and the local oscillator laser has to bephase locked to the incoming signal in order to achieve this. Forhomodyne detection the term e^(i(ω) ^(s) ^(−ω) ^(LO) ^()t+i(φ) ^(s)^((t)−φ) ^(LO) ^((t)) is) 1, and the beat term becomesRe└E_(s)(t)E_(LO)*┘

For the binary phase shift keying (BPSK) modulation format for example,E_(s)(t) takes on the value 1 or −1 depending on whether a logical “1”or “0” was transmitted, and the decision circuit can simply act on thebeat term directly.

With heterodyne detection there is a finite difference in opticalfrequency between the signal and local oscillator. All the amplitude andphase information on the signal appears on a carrier at angularfrequency (ω_(s)−ω_(LO)), the intermediate frequency, and it can bedetected with a demodulator using standard radio detection methods.Typically homodyne detection gives better performance than heterodynedetection, but is harder to implement because of the need for opticalphase locking. Heterodyne detection can be further divided into twocategories: synchronous and asynchronous. With synchronous heterodynedetection the receiver makes an estimate of the optical phase differencebetween the incoming optical signal and the light from the localoscillator, and applies the phase estimate during the digital decisionmaking process. An asynchronous heterodyne detection receiver does notmake an estimate of the phase. The data is obtained via another method,depending on which modulation format was used, such as by taking thedifference between one digital symbol and the next (differentialdetection). Synchronous detection gives better receiver sensitivity thanasynchronous detection. Homodyne detection can be considered to be asynchronous coherent detection method, because the process of opticalphase locking the local oscillator requires a phase estimate to be made.

C. Sampled Coherent Detection

A new method of coherent detection called sampled coherent detection hasbeen proposed and demonstrated recently, as described in U.S. PatentApplication No. 2004/0114939 and in “Coherent detection method using DSPfor demodulation of signal and subsequent equalization of propagationimpairments” by M. G. Taylor (IEEE Phot. Tech. Lett., vol. 16, no. 2, p.674-676, 2004), which are herein incorporated by reference. Digitalsignal processing (DSP) is employed in this method to obtain theinformation carried by a signal from the beat products seen at theoutputs of a phase diverse hybrid. The field of digital signalprocessing is summarized below.

In sampled coherent detection, the signal and local oscillator arecombined in a passive component called a phase and polarization diversehybrid. FIG. 2 shows a sampled coherent detection apparatus. The fouroutputs of the phase and polarization diverse hybrid are detected byseparate photodetectors 212 and then, after optional amplification byamplifiers 213, they are sampled by A/D converters 214. The samplevalues of the A/D converters are processed by the digital signalprocessor 215 to calculate the complex envelope of the signal electricfield over time. The phase and polarization diverse hybrid has fouroutputs 208-211 in the example of FIG. 2, where single ended detectionis used. The top two outputs 208 and 209 have the LO in one state ofpolarization, e.g., the horizontal polarization, and the lower twooutputs 210 and 211 have the LO in the orthogonal, vertical,polarization. For each of the two LO polarization states, the signal iscombined with the LO in a 900 hybrid 205, also known as a phase diversehybrid. The phase of the LO relative to the signal in one output of the90° hybrid is different by π/2 radians (i.e. 90°) compared to the phaseof the LO relative to the signal in the other output. This phase shiftcan be implemented by extra path length in one arm 207 of the 90° hybridcarrying the LO compared to the other arm 206, as can be seen in FIG. 2.The orthogonal SOP relationship between the two 90° hybrids is achievedby using a polarization beamsplitter 204 to divide light from the localoscillator 202 between the two hybrids and a standard 1:2 splitter 203to divide the incoming signal light 201.

The following mathematical treatment explains how the electric field ofthe signal is obtained from the outputs of the phase and polarizationdiverse hybrid. The incoming signal electric field can be written asRe└E_(s)(t)e^(iω) ^(s) ^(t+iφ) ^(s) ^((t))┘where E_(s)(t) is a Jones vector, a two-element vector comprising thepolarization components of the electric field in the horizontal andvertical directions. The use of Jones vectors is summarised in theAppendix. ${E_{s}(t)} = \begin{pmatrix}{E_{sx}(t)} \\{E_{sy}(t)}\end{pmatrix}$Each of the four outputs of the phase and polarization diverse hybrid inFIG. 2 contains signal Re└E_(s)(t)e^(iω) ^(s) ^(t+iφ) ^(s) ^((t))┘. Thelocal oscillator in the four outputs is different, and can be written asfollows

-   -   top output . . . Re└E_(LO)e^(iω) ^(LO) ^(t+iφ) ^(LO)        ^((t)){tilde over (x)}┘    -   2nd output . . . Re└i E_(LO)e^(iω) ^(LO) ^(t+iφ) ^(LO)        ^((t)){tilde over (y)}┘    -   3rd output . . . Re└E_(LO)e^(iω) ^(LO) ^(t+iφ) ^(LO)        ^((t)){tilde over (y)}┘    -   4th output . . . Re└i E_(LO)e^(iω) ^(LO) ^(t+iφ) ^(LO)        ^((t)){tilde over (y)}┘        In the top two arms the LO is horizontally polarized, in the        direction of Jones unit vector {tilde over (x)}, and in the        lower two arms vertical in the direction of {tilde over (y)}.        The π/2 phase shift is accounted for by the multiplicative        imaginary number i. The beat term parts of the optical powers in        the four outputs 208 through 211 are therefore        beat term 1=Re└E _(sx)(t)E _(LO) *e ^(i(ω) ^(s) ^(−ω) ^(LO)        ^()t+i(φ) ^(s) ^((t)−φ) ^(LO) ^((t))┘)        beat term 2=Im└E _(sx)(t)E _(LO) *e ^(i(ω) ^(s) ^(−ω) ^(LO)        ^()t+i(φ) ^(s) ^((t)−φ) ^(LO) ^((t))┘)        beat term 3=Re└E _(sy)(t)E _(LO) *e ^(i(ω) ^(s) ^(−ω) ^(LO)        ^()t+i(φ) ^(s) ^((t)−φ) ^(LO) ^((t))┘)        beat term 4=Im└E _(sy)(t)E _(LO) *e ^(i(ω) ^(s) ^(−ω) ^(LO)        ^()t+i(φ) ^(s) ^((t)−φ) ^(LO) ^((t))┘)        It follows that the envelope of the signal electric field is        related to the observed beat terms by $\begin{matrix}        {{E_{s}(t)} = {\frac{{\mathbb{e}}^{{{- {{\mathbb{i}}{({\omega_{s} - \omega_{LO}})}}}t} - {{\mathbb{i}}{({{\phi_{s}{(t)}} - {\phi_{LO}{(t)}}})}}}}{E_{LO}^{*}}\begin{pmatrix}        {\left( {{beat}\quad{term}\quad 1} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 2} \right)}} \\        {\left( {{beat}\quad{term}\quad 3} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 4} \right)}}        \end{pmatrix}}} & (2)        \end{matrix}$        In order to implement equation 2, the frequency difference        ω_(s)-ω_(LO) and phase difference φ_(s)(t)-φ_(LO)(t) must be        known. The digital signal processor must make estimates of these        quantities and apply these to make an estimate of the signal        electric field envelope. $\begin{matrix}        {{{\hat{E}}_{s}(t)} = {\frac{{\mathbb{e}}^{{{- {\mathbb{i}}}\hat{\omega}\quad t} - {{\mathbb{i}}{\hat{\phi}{(t)}}}}}{E_{LO}^{*}}\begin{pmatrix}        {\left( {{beat}\quad{term}\quad 1} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 2} \right)}} \\        {\left( {{beat}\quad{term}\quad 3} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 4} \right)}}        \end{pmatrix}}} & (3)        \end{matrix}$        Ê_(s)(t) is an estimate of the true signal electric field        envelope E_(s)(t), and the decision of the data carried by the        signal is derived from Ê_(s)(t). {circumflex over (ω)} is an        estimate of the angular frequency difference ω_(s)-ω_(LO).        {circumflex over (φ)}(t) is an estimate of the optical phase        difference between signal and local oscillator        φ_(s)(t)-φ_(LO)(t) . Clearly if the estimates {circumflex over        (ω)} and {circumflex over (φ)}(t) are exactly correct, then        equation 3 becomes equation 2, and the estimate of the signal        electric field envelope Ê_(s)(t) is equal to the true envelope        E_(s)(t). Conversely, any inaccuracy in the estimates        {circumflex over (ω)} and {circumflex over (φ)}(t) leads to        errors in the received digital data. The object of the present        invention is to make an accurate phase estimate within the        digital signal processor.

The digital information is imposed on the signal electric field envelopeas a sequence of symbols regularly spaced in time. To recover theinformation the digital signal processor must also have an estimate ofthe times of the symbol centers (i.e., a symbol clock).

Transmission over a length of optical fiber transforms the state ofpolarization of an optical signal, so that the digital values taken onby E_(s)(t) as seen at the receive end of a fiber optic transmissionsystem are typically not the same as those imposed at the transmit end.The polarization transformation can be reversed within the DSP bymultiplying by the appropriate rotation Jones matrix R, so that thefirst element of the Jones vector contains the complex envelope of aninformation-bearing signal. $\begin{matrix}{{{\hat{E}}_{s}(t)} = {\frac{{\mathbb{e}}^{{{- {\mathbb{i}}}\hat{\omega}\quad t} - {{\mathbb{i}}{\hat{\phi}{(t)}}}}}{E_{LO}^{*}}{\overset{\sim}{x} \cdot {R\begin{pmatrix}{\left( {{beat}\quad{term}\quad 1} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 2} \right)}} \\{\left( {{beat}\quad{term}\quad 3} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 4} \right)}}\end{pmatrix}}}}} & (4)\end{matrix}$(Ê_(s)(t) is written without boldface because it represents a complexelectric field without regard to polarization, and not a Jones vector.)The correct rotation matrix R can be estimated by exploring theavailable space and then locking on to the matrix which gives the bestquality signal. The polarization transformation of the optical fibertypically changes slowly, so the rotation matrix must be allowed toupdate. Alternatively the SOP of the local oscillator can be matched tothat of the signal by a hardware polarization controller, so thatequation 3 need be implemented only for one element of the Jones vectorE_(s)(t), based on two phase diverse hybrid outputs instead of four.$\begin{matrix}{{{\hat{E}}_{s}(t)} = {\frac{{\mathbb{e}}^{{{- {\mathbb{i}}}\hat{\omega}\quad t} - {{\mathbb{i}}{\hat{\phi}{(t)}}}}}{E_{LO}^{*}}\left( {\left( {{beat}\quad{term}\quad 1} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 2} \right)}} \right)}} & (5)\end{matrix}$

The Jones vector E_(s)(t) constitutes a complete description of theoptical signal, or more precisely of the signal's optical spectrum inthe region of the local oscillator. This means that any parameter of theoptical signal can be deduced from E_(s)(t). Employing sampled coherentdetection is more complex than direct detection, but has many benefits.Phase encoded modulation formats can be employed, such as BPSK andquadrature phase shift keying (QPSK), which offer better sensitivitythan on-off modulation formats. Also polarization multiplexed formatscan be employed, which offer twice the information capacity for a givenbandwidth of electro-optic components and a given optical spectralbandwidth. The polarization demultiplexing operation is performed withinthe digital signal processor, so no additional optical components areneeded for it. In a long fiber optic transmission system carrying highbit rate signals, the optical fiber propagation effects, such aschromatic dispersion and polarization mode dispersion, distort thesignals. With sampled coherent detection, the propagation effects can bereversed within the DSP by applying an appropriate mathematicaloperation. Finally, a key benefit of sampled coherent detection is thatit is equivalent to passing the signal through a narrow optical filtercentred on the local oscillator wavelength, so no narrow optical filtercomponents are needed for WDM. The LO can be tuned in wavelength, whichis equivalent to tuning the optical filter.

D. Digital Signal Processing

The present invention utilizes digital signal processing (DSP). DSP isdescribed in “Understanding Digital Signal Processing” by R. G. Lyons(Prentice Hall, 1996) and “Digital Signal Processing: Principles,Algorithms and Applications” by J. G. Proakis & D. Manolakis (PrenticeHall, 3rd ed., 1995), herein incorporated by reference. A signalprocessor is a unit which takes in a signal, typically a voltage vs.time, and performs a predictable transformation on it, which can bedescribed by a mathematical function. FIG. 3 a shows a generic analogsignal processor (ASP). The box 302 transforms the input signal voltage301 into the output signal voltage 303, and may contain a circuit ofcapacitors, resistors, inductors, transistors, etc. FIG. 3 b illustratesa digital signal processor. First, the input signal 301 is digitized bythe analog to digital (A/D) converter 304, that is converted into asequence of numbers, each number representing a discrete time sample.The core processor 305 uses the input numerical values to compute therequired output numerical values, according to a mathematical formulathat produces the required signal processing behavior. The output valuesare then converted into a continuous voltage vs. time by the digital toanalog (D/A) converter 306. Alternatively, for applications in a digitalsignal receiver, the analog output of the DSP may go into a decisioncircuit to produce a digital output. In such a situation, the digitalprocessing core may perform the decision operation and output theresult, in which case the D/A converter 306 is not needed.

The digital filter is an operation that may be executed in a digitalsignal processor. A sequence of filtered output values Y(n), n=0, 1, 2,3 . . . , is calculated from input values X(n) by $\begin{matrix}{{Y(n)} = {{\sum\limits_{k = 0}^{B}{{b(k)}{X\left( {n - k} \right)}}} + {\sum\limits_{k = 1}^{A}{{a(k)}{Y\left( {n - k} \right)}}}}} & (6)\end{matrix}$The b(k) are known as feedforward tap weights, and the a(k) are feedbacktap weights. The digital filter of equation 6 may be written in terms ofa z-transfer function${Y(z)} = {\frac{\sum\limits_{k = 0}^{B}{{b(k)}z^{- k}}}{1 - {\sum\limits_{k = 1}^{A}{{a(k)}z^{- k}}}} - {X(z)}}$X(z) and Y(z) are the z-transforms of X(n) and Y(n) respectively.

The applications of digital signal processing in optical coherentdetection of communications signals may be such that the clock speed ofthe logic in the digital signal processor is slower than the symbol rateat which information is transmitted. For example, the DSP clock speedmay be 1 GHz while the information signaling rate is 10 Gbaud. Thismeans that the DSP must operate in parallel. The incoming symbols arefirst demultiplexed into parallel branches and then each branch isprocessed at the low logic clock speed. A constraint of such a parallelarchitecture is that symbol n-1 may be processed at the same time assymbol n, so the result of an operation on symbol n-1 is not availableat the commencement of the operation on symbol n. Many signal processingalgorithms feed back the result of operation n-1 to calculate operationn. Such an algorithm cannot be implemented in a parallel digitalprocessor. The parallel operation of the DSP is equivalent to imposing adelay on any feedback paths. It is an object of the present invention toestimate the optical phase in the digital signal processor without usingfeedback from immediately preceding results, but instead employingalgorithms that do not use any feedback or that use feedback fromdistant past results.

E. Existing Phase Estimation Methods

Digital communications over radio and electrical cables has led to thedevelopment of methods for estimation of the phase of the carrier of anarrowband signal. Many of these methods are described in “DigitalCommunications” by J. G. Proakis (McGraw-Hill, 4th ed., 2000), “Digitalcommunication receivers: synchronization, channel estimation & signalprocessing” by H. Meyr, M. Moeneclaey & S. A. Fechtel (Wiley, 1998) and“Synchronization techniques for digital receivers” by U. Mengali & A. N.D'Andrea (Plenum Press, 1997), which are herein incorporated byreference. The delay to the times of the symbol centers, the frequencydifference between the signal and local oscillator and the phase of thesignal compared to the local oscillator are collectively known asreference parameters or as synchronization parameters. With coherentoptical detection there is an additional reference parameter: the stateof polarization of the signal compared to the LO. The best possibleestimate of a reference parameter that can be made based on the noisyobservations is known as the optimal estimate. If a reference parameterchanges unpredictably, but these changes are slow, then it is possibleto use an estimate which deviates from the optimal estimate withoutcausing a significant increase in the number of errors in the detecteddigital information. Also an algorithm with feedback to distant pastresults may be used. This means that a simple algorithm convenient forimplementation in the DSP may be chosen to estimate a slowly varyingreference parameter. However, it is important to make an estimate closeto the optimal estimate for a reference parameter that changes rapidlyon the time scale of the symbol period, otherwise there will be asubstantial increase in the bit error rate. In a typical implementationof coherent detection of optical signals, the symbol clock, thefrequency difference between signal and local oscillator, and the SOP ofthe signal all vary slowly. Techniques are available to estimate theseparameters. However, the optical phase of the signal compared to thelocal oscillator may vary rapidly and randomly, unless expensive narrowlinewidth lasers are used. A near optimal estimate of the phase istherefore needed. It is clear that any phase estimation method from thefield of radio can be applied to optical coherent detection. Howeverthere is no application in radio with the same level of phase noise aswith optical coherent detection. In a possible configuration usingdecision feedback (DFB) lasers, the combined laser linewidth Δν may be10 MHz and the signaling rate 10 Gbaud, so that the product of symboltime and linewidth is τ_(s)Δν=10⁻³. There are no examples in the priorart having such a high τ_(s)Δv, and the methods used in radio cannot beapplied to optical coherent detection.

Experiments were performed with synchronous optical coherent detectionin the early 1990's which inherently made an estimate of the phase.Examples are “4-Gb/s PSK Homodyne Transmission System Using Phase-LockedSemiconductor Lasers” by J. M. Kahn et al. (IEEE Phot. Tech. Lett., vol.2, no. 4, p. 285-287, 1990) and “An 8 Gb/s QPSK Optical HomodyneDetection Experiment Using External-Cavity Laser Diodes” by S. Norimatsuet al. (IEEE Phot. Tech. Lett., vol. 4, no. 7, p. 765-767, 1992). Theseexperiments used continuous time analog signal processing, but it isclear that an equivalent discrete time algorithm could be derived fromthe analog signal processing function. The experiments used decisiondirected detection and phase locked loops, both of which use feedback.In the experiments the feedback paths were kept purposely short. It wasknown that long feedback delay times imposed a requirement for narrowlinewidth lasers, as is discussed in “Damping factor influence onlinewidth requirements for optical PSK coherent detection systems” by S.Norimatsu & K. Iwashita (IEEE J. Lightwave Technol., vol. 11, no. 7, p.1226-1233, 1993). In a DSP implementation the effective length of thefeedback path is constrained by the clock frequency of the DSP logic.Also the experiments used external cavity lasers having low linewidthinstead of less expensive integrated semiconductor lasers, such as DFBlasers. So the techniques used in these optical coherent detectionexperiments do not provide a solution for implementing optical coherentdetection using digital signal processing with inexpensive widelinewidth lasers.

“PLL-Free Synchronous QPSK Polarization Multiplex/Diversity ReceiverConcept With Digital I&Q Baseband Processing” by R. Noé (IEEE Phot.Tech. Lett., vol. 17, no. 4, p. 887-889, 2005) discloses a phaseestimation method for optical coherent detection using digital signalprocessing, which is suitable for a parallel DSP because it does notemploy feedback. The method has been implemented experimentally in“Unrepeatered optical transmission of 20 Gbit/s quadrature phase-shiftkeying signals over 210 km using homodyne phase-diversity receiver anddigital signal processing” by D.-S. Ly-Gagnon et al. (IEE Electron.Lett., vol. 41, no. 4, p. 59-60, 2005). This phase estimation methodinvolves applying a fourth power nonlinearity and then taking a simpleaverage of complex field values of a group of contiguous symbols, whichis equivalent to planar filtering with a rectangular time responsefilter. The rectangular time response filter is chosen because it doesnot employ feedback. However, this filter shape is not close to theoptimal filter shape, which increases the bit error rate. Differentiallogical detection is used after decisions are made to avoid the impactof cycle slips, but it has the disadvantage that it increases the biterror rate. Another disadvantage is that every time the filter complexoutput crosses the negative real boundary (every time a cycle countoccurs) an extra symbol error is inserted, which leads to a backgroundbit error rate even when the transmission system additive noise is low.It would be better to include a cycle count function to avoid the effectof cycle slips, but cycle count functions use feedback.

Although not proposed for applications like optical coherent detectionwhere the oscillators have high linewidth, there are phase estimationsolutions in the field of radio that work with a randomly varying phase.A method is disclosed in “Digital communication receivers:synchronization, channel estimation & signal processing” by H. Meyr etal. where the complex field is operated on by a nonlinear function andthen passed to a planar filter having transfer function${H(z)} = \frac{1 - \alpha_{pp}}{1 - {\alpha_{pp}z^{- 1}}}$This is similar to the zero lag Wiener filter phase estimation method ofthe present invention which is described below, and has the transferfunction of equation 13. However the disclosure does not identify how tochoose α_(app) to obtain a close approximation to the optimal zero-lagphase estimate. Also the transfer function involves feedback to theimmediately preceding result, and the disclosure does not explain how toimplement the filter in a parallel digital signal processor. The samedisclosure describes how to unwrap the phase of the planar filter outputby using a cycle count function. The basic equation of the phaseunwrapping method is the same as equation 22 that is used in the presentinvention, described below. However the cycle count function in thedisclosure (and that of equation 22) employs feedback from theimmediately preceding result, and the disclosure does not explain how toimplement the cycle count function in a parallel digital signalprocessor.

Thus there is a need for a phase estimation method that can beimplemented in a parallel digital signal processor architecture, andwhich does not use feedback from recent results. There is also a furtherneed for such a phase estimation method to provide an estimate which isclose to the optimal phase estimate. It is an object of the presentinvention to provide such a methodology and system.

SUMMARY OF THE INVENTION

In one aspect, the present invention is a method and apparatus to makean estimate of the phase of a signal relative to the local oscillator inan optical coherent detection subsystem that employs a digital signalprocessor having a parallel architecture. The phase estimation methodcomprises operations that do not use feedback of recent results. Themethod includes a cycle count function so that the phase estimate leadsto few cycle slips. The phase estimate of the present invention isapproximately the same as the optimal phase estimate.

The digital signal processor in the optical coherent detection apparatusforms a sequence of complex values corresponding to the electric fieldof the symbols before a phase estimate is applied. The method of makinga phase estimate has the following stages. First, a nonlinear functionis applied to the sequence of electric field values to remove theaveraging effect of the transmitted data. Then a digital filter isapplied, whose transfer function is a Wiener filter. The feedback andfeedforward taps of the filter are chosen using a look-aheadcomputation, so that the filter has the correct Wiener filter transferfunction, but it can be implemented without feedback of recent results.The Wiener filter is the optimal linear estimate of the phase. Two kindsof Wiener filter may be used, zero-lag or finite-lag filters. Next acycle count operation is performed. The cycle count function is derivedusing a look-ahead computation, so that it can be implemented withoutfeedback of recent results. The nonlinear function is then reversed togive a sequence of complex values whose phases are the final phaseestimate. The reversal of the nonlinear function is performed takinginto account the cycle count. This means that there are few cycle slips.After applying the phase factor to the original sequence of electricfield values, a decision is made on the digital value of each symbol.Differential logical detection may be subsequently applied to remove theimpact of any remaining cycle slips. This may be followed by decoding ofany forward error correction code on the transmitted data. The FEC codemay be chosen to correct short bursts of errors, so that thedifferential logical detection operation does not lead to excess biterrors.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 (Prior Art) illustrates an existing coherent receiver;

FIG. 2 (Prior Art) illustrates a polarization and phase diverseconfiguration for sampled coherent detection;

FIG. 3 a (Prior Art) illustrates a generic diagram of an analog signalprocessor;

FIG. 3 b (Prior Art) illustrates a generic diagram of a digital signalprocessor;

FIG. 4 illustrates the phase estimation process in the preferredembodiment of the present invention;

FIG. 5 a illustrates an example of the impulse response function of azero-lag Wiener filter;

FIG. 5 b illustrates an example of the impulse response function of afinite-lag Wiener filter;

FIG. 6 illustrates an example of an approximate Wiener filter impulseresponse; and

FIG. 7 illustrates the process of obtaining the information contained onan optical signal by coherent detection, followed by phase estimationand differential logical detection in the preferred embodiment of thepresent invention.

DESCRIPTION OF THE INVENTION

The present invention is a phase estimation method that can beimplemented in a parallel digital signal processor architecture, andwhich does not use feedback from recent results. There are manymodulation formats used to transmit digital information on subcarriers.The methods used to estimate the phase may differ in the details, buthave a common basic approach, as is discussed in “Digital communicationreceivers: synchronization, channel estimation & signal processing” byH. Meyr et al. and “Synchronization techniques for digital receivers” byU. Mengali & A. N. D'Andrea. The method to estimate the phase of an MaryPSK signal will be described below. It will be apparent to those skilledin the art that the present invention for making phase estimates may beutilized for other modulation formats, such as offset QPSK or QAM.

FIG. 4 illustrates the phase estimation process in the preferredembodiment of the present invention. The optical signal 401 is input toa sampled coherent detection unit 402. The frequency difference betweensignal and local oscillator 403 and the state of polarization of thesignal compared to the local oscillator may be estimated accurately byknown methods, since these reference parameters vary slowly in a typicaloptical transmission system. The complex variable r(t) is formed byapplying compensations for the frequency difference a and for the signalSOP to the electric field envelope derived from the beat terms in theoptical powers at the phase diverse hybrid outputs. In the case of thepolarization diverse coherent detection configuration, in analogy withequation 4${r(t)} = {\frac{{\mathbb{e}}^{{- {\mathbb{i}}}\hat{\omega}\quad t}}{E_{LO}^{*}}{\overset{\sim}{x} \cdot {R\begin{pmatrix}{\left( {{beat}\quad{term}\quad 1} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 2} \right)}} \\{\left( {{beat}\quad{term}\quad 3} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 4} \right)}}\end{pmatrix}}}}$Similarly for the case where the SOP of the local oscillator is alignedwith that of the signal by a hardware polarization controller, inanalogy with equation 5${r(t)} = {\frac{{\mathbb{e}}^{{- {\mathbb{i}}}\hat{\omega}\quad t}}{E_{LO}^{*}}\left( {\left( {{beat}\quad{term}\quad 1} \right) + {{\mathbb{i}}\left( {{beat}\quad{term}\quad 2} \right)}} \right)}$An alternative to compensating for the frequency difference {circumflexover (ω)} within the DSP is to control the optical frequency of thelocal oscillator laser so that the frequency difference is zero. r(t) isrelated to the signal envelope of the transmitted optical signal (or toone of the transmitted polarization multiplexed tributaries in the caseof a polarization multiplexed optical signal) byr(t)=e^(iφ(t)) E _(s)(t)   (7)whereφ(t)=φ_(s)(t)−φ_(LO)(t)The task is to determine φ(t) from the available values of r(t), so thatE_(s)(t) can be calculated by inverting equation 7.

The symbol clock may also be determined by existing techniques. Thedigital signal processor is able to calculate the values of r(t) at thecenters of the symbolsr(nτ _(s)) n=0,1,2,3 . . .where it is assumed that the origin of time t is chosen so as to lie ata symbol center. In this document r(nτ_(s)) will be written as r(n) forcompactness. Any variable which is a function of t is understood to be acontinuous time function; a variable which is a function of n representsthe equivalent discrete time function sampled at the symbol centers; anda variable which is a function of z represents the z-transform of thediscrete-time variable. The SOP estimation, frequency differencecompensation and the process or retiming according to the recoveredsymbol clock are performed in element 404 of FIG. 4 within the digitalsignal processor 410.

If the transmitter and local oscillator lasers have Lorentzianlineshape, which is the typical laser lineshape, thenφ(n)=φ(n−1)+w(n)   (8)where w(n) is a zero-mean Gaussian noise sequence. The variance σ_(w) ²of w(n) is related to the combined full width half maximum linewidth Δvof the lasers byσ_(w) ²=2πτ₂Δν

The electric field envelope imposed on the transmitted signal by datamodulation isE _(s)(n)=d(n)E _(s0)   (9)where E_(s0) is a real constant and d(n) is a sequence of values codingthe data to be transmitted, these values being taken from the M-memberset M√{square root over (1)}. For BPSK the set is {−1,1}, and for QPSKit is {−i,−1,i,1}. After modulation onto the transmitter laser,transmission through a fiber optic transmission system and detection,the received signal is (ignoring the scaling factor experienced byE_(s0) from transmitter to receiver)r(n)=E _(s0) e ^(iφ(n))(d(n)+p(n))   (10)where p(n) is a zero-mean complex Gaussian noise sequence whose real andimaginary parts each have variance σ_(p) ², and which represents theadditive noise whose variance (of each complex component) is E_(s0)²σ_(p) ².

The next step in the phase estimation process executed in element 405 bythe digital signal processor is to apply a power law nonlinear functionto the observed values r(n)s(n)=r(n)^(M)From equation 10, and since d(n)^(M)=1 for all possible values of d(n),it follows thats(n)=E _(s0) ^(M) e ^(iMφ(n))(1+Mp′(n))   (11)where p′(n) is a different zero-mean complex Gaussian noise sequencefrom p(n), but which also has real and imaginary parts with varianceσ_(p) ²; and terms of order p(n)² or higher have been neglected giventhat the signal-to-noise ratio (SNR) is large. Taking the phase angle ofboth sides of equation 11, and assuming that a small angle approximationcan be applied given that the SNR is largeθ(n)=Mφ(n)+M Im[p′(n)]  (12)whereθ(n)=arg _(unwrapped) [s(n)]θ(n) and φ(n) are both unwrapped phase angles (i.e., lie in the range−∞) to ∞).

Studying equation 12, which is approximately true when the SNR is high,Im[p′(n)] is a zero-mean real Gaussian noise sequence having varianceσ_(p) ², and the statistics of φ(n) are known from equation 8. Accordingto estimation theory, the best linear estimate of φ(n) is made byapplying a Wiener filter to the observable quantity θ(n). There are twokinds of Wiener filter that can be applied: a filter having zero lag anda filter having a lag of D symbols. Following the prescription laid outin “Digital Signal Processing: Principles, Algorithms and Applications”by J. G. Proakis & D. Manolakis, the zero-lag Wiener filter has transferfunction $\begin{matrix}{{\hat{\phi}(z)} = {\frac{1}{M}\frac{1 - \alpha}{1 - {\alpha\quad z^{- 1}}}{\theta(z)}}} & (13)\end{matrix}$and the finite-lag filter has transfer function $\begin{matrix}{{{\hat{\phi}(z)} = {\frac{1}{M}\frac{{\left( {1 - \alpha} \right)\alpha^{D}} + {\left( {1 - \alpha} \right)^{2}{\sum\limits_{k = 1}^{D}{\alpha^{D - k}z^{- k}}}}}{1 - {\alpha\quad z^{- 1}}}{\theta(z)}}}{where}} & (14) \\{\alpha = \frac{\sigma_{w}^{2} + {2\sigma_{p}^{2}} - {\sigma_{w}\sqrt{\sigma_{w}^{2} + {4\sigma_{p}^{2}}}}}{2\sigma_{p}^{2}}} & (15)\end{matrix}$The role of the Wiener filter is to smooth out the rapidly varyingrandom contribution from the additive noise to obtain a result which isclose to the inertial randomly changing phase. The finite-lag filterworks better than the zero-lag filter because it effectively looksforward in time by D symbols to decide on its smoothed output, as wellas looking at the infinite past. The Wiener filter transfer functions ofequations 13 and 14 have been derived assuming the phase noise followsthe Gaussian random walk statistics of equation 8 and the additive noiseis Gaussian. The appropriate Wiener filter may be calculated if the twonoise sources have other statistical behavior.

The digital filter is implemented with the feedforward taps determinedby the coefficients of the powers of z in the numerator, and thefeedback taps determined by the coefficients in the denominator,according to the well known principles of digital signal processing. TheWiener filters cannot be directly realized in a parallel digital signalprocessor, however, because the denominators of both equations 13 and 14contain terms in z⁻¹, indicating feedback of the immediately precedingresult. To resolve this issue, the Wiener filter algorithms may berecast using a look-ahead computation, as is described in “VLSI digitalsignal processing systems: Design and implementation” by K. K. Parhi(Wiley, 1999), which is herein incorporated by reference. The look-aheadcomputation involves replacing all terms in an algorithm of result n-Cwith the expression containing terms n-C-1 and older, and repeating thisprocess until the algorithm contains only sufficiently old results thatit can be implemented. For the z-transfer functions of equations 13 and14, recasting using a look-ahead computation is expressed as multiplyingboth numerator and denominator by the same polynomial$\sum\limits_{k = 0}^{L - 1}{\alpha^{k}z^{- k}}$where L is a suitably chosen positive integer. Equation 13 for thezero-lag Wiener filter becomes $\begin{matrix}{{\hat{\phi}(z)} = {\frac{1}{M}\frac{\left( {1 - \alpha} \right){\sum\limits_{k = 1}^{L - 1}{\alpha^{k}z^{- k}}}}{1 - {\alpha^{L}z^{- L}}}{\theta(z)}}} & (16)\end{matrix}$and equation 14 for the finite-lag Wiener filter becomes $\begin{matrix}{{\hat{\phi}(z)} = {\frac{1}{M}\frac{\left( {{\left( {1 - \alpha} \right)\alpha^{D}} + {\left( {1 - \alpha} \right)^{2}{\sum\limits_{k = 1}^{D}{\alpha^{D - k}z^{- k}}}}} \right){\sum\limits_{k = 1}^{L - 1}{\alpha^{k}z^{- k}}}}{1 - {\alpha^{L}z^{- L}}}{\theta(z)}}} & (17)\end{matrix}$Both z-transfer functions now have a denominator containing z^(−L),which refers to feedback of a result L symbols in the past, where L canbe chosen by the DSP designer. Hence, both Wiener filters may now beimplemented directly in a parallel digital signal processor. Thenumerator in equation 16 is written as the product of two sums, and theproduct may be expanded to give the feedforward tap weights for a givenD and L. The digital filters have become more complicated, in that thenumber of feedforward taps has increased from D+1 to D+L. If α^(L) isvery small then the feedback tap can be omitted, and the digital filterbecomes a feedforward-only filter.

There are several ways to implement the digital filter, by acting ondifferent quantities. The phase estimate may take the form of a phaseangle {circumflex over (φ)}(n) or a phase factore^(i{circumflex over (φ)}(n)). In one embodiment of the presentinvention, the digital signal processor calculates the phase angle ofvalues s(n) using a look-up table, for example, unwraps the phase usingthe method described below, and then applies the Wiener filter to thevalues of θ(n) to obtain the phase estimate {circumflex over (φ)}(n). Ina second preferred embodiment of the invention, the Wiener filter isapplied to the complex values s(n) as a planar filter, instead of to thephase angle values θ(n). A parameter u(z) is calculated in element 406in FIG. 4 by applying the digital filter to s(n). In the case of thezero-lag Wiener filter $\begin{matrix}{{u(z)} = {\frac{\left( {1 - \alpha} \right){\sum\limits_{k = 1}^{L - 1}{\alpha^{k}z^{- k}}}}{1 - {\alpha^{L}z^{- L}}}{s(z)}}} & (18)\end{matrix}$and for the finite-lag Wiener filter $\begin{matrix}{{u(z)} = {\frac{1}{M}\frac{\left( {{\left( {1 - \alpha} \right)\alpha^{D}} + {\left( {1 - \alpha} \right)^{2}{\sum\limits_{k = 1}^{D}{\alpha^{D - k}z^{- k}}}}} \right){\sum\limits_{k = 1}^{L - 1}{\alpha^{k}z^{- k}}}}{1 - {\alpha^{L}z^{- L}}}{s(z)}}} & (19)\end{matrix}$It can be verified that the phase angle of u(z) is approximately equalto M{circumflex over (φ)}(n) given that a small angle approximationapplies. The phase factor may therefore be calculated in element 408$\begin{matrix}{{\mathbb{e}}^{{- {\mathbb{i}}}{\hat{\phi}{(n)}}} = \sqrt[M]{\frac{u^{*}(n)}{{u(n)}}}} & (20)\end{matrix}$and this phase factor 409 is applied to calculate the required electricfield envelope Ê_(s)(n)Ê _(s)(n)=e ^(−i{circumflex over (φ)}(n)) r(n)   (21)The data estimate {circumflex over (d)}(n) may then be obtained viaequation 9. The Mth root in equation 20 may take on M possible values,and the root must be chosen taking into account the cycle count of u(n),as is discussed below. Simulations indicate that for a given additivenoise and combined laser linewidth, the preferred embodiment of theinvention using planar filtering (using equations 18, 19 and 20) leadsto a lower bit error rate than the alternative embodiment where thedigital filter acts on the phase angle of s(n).

Both of the embodiments described above make use of a cycle countfunction. The cycle count of a complex variable is an integer which isincremented every time the variable crosses the negative real axis in anincreasing phase direction, and decremented every time the variablecrosses that axis in a decreasing phase direction. The cycle countc_(u)(n)of complex variable u(n) may be calculated in element 407 asfollowsc _(u)(n)=c _(u)(n−1)+ƒ(u(n−1),u(n))   (22)whereƒ(X,Y)=−1 arg[Y]−arg[X]>πƒ(X,Y)=0 |arg[Y]−arg[X]|≦πƒ(X,Y)=1 arg[Y]−arg[X]<−πarg[•] is the unwrapped phase angle lying between −π and π. Anequivalent expression for the function ƒ(X,Y) which may be used isƒ(X,Y)=−1 Im[Y]≧0, Im[X]<0, Im└X*Y┘<0ƒ(X,Y)=1 Im[Y]<0, Im[X]≧0, Im└X*Y┘>0

-   -   ƒ(X,Y)=0 otherwise        The cycle count expression of equation 22 uses only u(n−1) and        u(n) to make each cycle count decision. More complicated cycle        count functions are possible using more samples, which are less        likely to give an incorrect estimate corresponding to a cycle        slip. The cycle count need only be recorded modulo M. The        problem of choosing which Mth root in equation 20 is resolved        using the cycle count function. $\begin{matrix}        {{\mathbb{e}}^{{- {\mathbb{i}}}{\hat{\phi}{(n)}}} = {\sqrt[M]{\frac{u^{*}(n)}{{u(n)}}}{\mathbb{e}}^{{- {{\mathbb{i}c}_{u}{(n)}}}2{\pi/M}}}} & (23)        \end{matrix}$        where the Mth root in equation 23 refers to the principal Mth        root. The alternative embodiment described above made use of        θ(n), the unwrapped phase angle of s(n). The unwrapped phase        angle may be calculated as follows        θ(n)=arg _(unwrapped) [s(n)]=arg[s(n)]+2π c _(s)(n)        where c_(s)(n) is the cycle count of s(n) in analogy with        equation 22.

The cycle count method of equation 22 cannot be implemented directly ina parallel digital signal processor, however, because it uses feedbackfrom result c_(u)(n−1) to calculate c_(u)(n). This issue can be resolvedby recasting equation 22 using a look-ahead computation. $\begin{matrix}{{c_{u}(n)} = {{c_{u}\left( {n - L} \right)} + {\sum\limits_{k = 0}^{L - 1}{f\left( {{u\left( {n - k - 1} \right)},{u\left( {n - k} \right)}} \right)}}}} & (24)\end{matrix}$L may be chosen by the DSP designer to be large enough to avoid feedbackoperations that are not allowed. Equation 24 may be implemented in aparallel digital processor, although it is more complicated than theoriginal algorithm of equation 22.

The parameter α appearing in equations 16 and 17 must be estimatedaccurately in order to use the correct Wiener filter. α depends on σ_(w)and σ_(p), according to equation 15. Before the phase estimate has beenapplied, σ_(p) may be estimated from |r(n)|², since${2\sigma_{p}^{2}} = {\frac{\overset{\_}{{r}^{2}}}{\sqrt{{2\left( \overset{\_}{{r}^{2}} \right)^{2}} - \overset{\_}{{r}^{4}}}} - 1}$The bar over a quantity indicates the time average. σ_(p) may beestimated based on the time average of a moderate number of symbols.Similarly, σ_(w) may be estimated fromM ² Kσ _(w) ²=variance(arg[(r(n−K)*r(n))^(M)])−2M ²σ_(p) ²where constant K is chosen large enough that the phase noise dominatesover the effect of additive noise, but not so large that cycle slipsoccur, making the estimate inaccurate. Once phase estimation has startedbased on a preliminary value of σ_(w), a better estimate of σ_(w) may bemade using a decision directed estimate.

The use of a digital filter having the Wiener filter response leads to aphase estimate which is close to the optimum phase estimate. Clearly isalso possible to employ a digital filter which is an approximation ofthe Wiener filter. The phase estimate will then be less accurate, but itmay be acceptable. The approximate Wiener filter may be easier toimplement than the exact Wiener filter. Following is an a example of anapproximate Wiener filter. The implementation of the Wiener filterindicated by either of equations 16 and 17 has many feedforward taps,each with a different weight. FIG. 5 a illustrates an example of themagnitude of the impulse response function of a zero-lag Wiener filter.FIG. 5 b illustrates an example of the magnitude of the impulse responsefunction of a finite-lag Wiener filter. The time response functionsshown in FIG. 5 are examples of the feedforward tap weights for large Lfor the zero-lag Wiener filter 502 and the finite-lag Wiener filter 502.The multiplication operation requires more resources in a digital signalprocessor than the addition operation. Therefore it is easier toapproximate the smoothly varying weights by a step-changing function.Such a step changing function is illustrated in FIG. 6, where 601 is anapproximation of the finite-lag Wiener filter feedforward tap weights.The DSP may then sum groups of contiguous samples of u(n), and apply adigital filter such that each sum corresponds to one digital filter tap.

The present invention includes digital filters that approximate theWiener filter response. It is necessary to state how close the digitalfilter must be to the Wiener filter in order to distinguish the presentinvention from the prior art, for example the planar filter withrectangular impulse response of “PLL-Free Synchronous QPSK PolarizationMultiplex/Diversity Receiver Concept With Digital I&Q BasebandProcessing” by R. Noé. Any linear filter is described by its impulseresponse (and a nonlinear filter may be described by the impulseresponse if its linear part). The time scale of the impulse response maybe chosen so that the origin (n=0 point) corresponds to the point atwhich the estimate is made. Thus, a finite-lag filter has some non-zerovalues of impulse response for n>0 in addition to negative time, while azero-lag filter has only zero time and negative time response. Thepresent invention includes a digital filter which has an impulseresponse whose non-zero values have the trend of decaying for increasing|n|. When the filter is a finite-lag filter the impulse responsetherefore has a decaying trend for both the positive and negative timedirections. When the filter is a zero-lag filter the trend is for decayin negative time. In addition, as has already been discussed above, thepresent invention uses either no feedback or it uses feedback of distantpast results.

Although the number of cycle slips is low with an accurate cycle countincluded in the phase estimate, it is possible that the received symbolsequence Ê_(s)(n) contains some cycle slips compared to the transmittedsymbol sequence E_(s)(n). The effect of a cycle slip at symbol n is aphase rotation of that symbol and all subsequent symbols, so that allreceived data following symbol n may be incorrect. In practice, the higherror rate will trigger a reset of the receiver, and the phase will becorrected at a subsequent framing word in the transmitted symbolsequence. The overall impact of the cycle slip may then be a largenumber of contiguous symbol errors, between the occurrence of the cycleslip and the framing word when the phase was reset. The number oferrored symbols may be too large to be corrected by forward errorcorrection decoding, so that the end recipient of the informationexperiences a large number of errors. The probability of cycle slip mustbe engineered to be much lower than the probability of bit error in thetransmission system (before FEC correction). The number of cycle slipsscales negative exponentially with the square root of the combinedlinewidth. Low cycle slip probability may be achieved by using narrowlinewidth lasers, but this is undesirable because of the expense of thelasers. Simulations indicate that the linewidth requirement on thelasers to achieve a cycle slip probability sufficiently low fortelecommunications applications is more stringent than the linewidthrequirement to achieve a low bit error rate increase, by about twoorders of magnitude.

FIG. 7 illustrates the DSP 410 multiplying the output of the phaseestimator 701 by r(n) at multiplier 702 to give Ê_(s)(n), in accordancewith equation 21. A decision is made at element 703 of the data symbol{circumflex over (d)}(n). The persistent nature of the impact of a cycleslip may be avoided by using differential logical detection (also knownin the field of radio as differentially coherent detection) 704 afterthe symbol decision, and applying the appropriate preceding at thetransmitter, as is described in “Synchronization techniques for digitalreceivers” by U. Mengali & A. N. D'Andrea. With differential logicaldetection the information content of the optical signal (before FECdecoding) is obtained by comparing the value of one symbol to an earliersymbol, usually the immediately preceding symbol. The information is thesequence {circumflex over (d)}(n−1)*{circumflex over (d)}(n). Afterdifferential logical detection, a single cycle slip event becomes asingle isolated symbol error. A disadvantage of differential logicaldetection is that each symbol error before differential logicaldetection is converted to a pair of contiguous symbol errors, equivalentto a short burst of bit errors. The increase in bit error rate reducesthe advantage of using synchronous coherent detection over asynchronouscoherent detection.

The output of the differential logical detection operation is decoded byan FEC decoder 705 to give the final data output 706. The increase inchannel bit error rate due to differential logical detection may bereversed by employing an FEC code which inherently corrects short burstsof errors. There are many codes to choose from which inherently correctshort error bursts, for example Reed-Solomon codes or cyclic codes. Thestatement that a block FEC code inherently corrects short bursts oferrors means the following. A case of maximally errored block may beidentified where a received block contains the maximum number of biterrors for complete error correction. After decoding, such a blockcontains no bit errors. The addition of one more isolated bit error tothe maximally errored block will cause the block to contain one or morebit errors after decoding. However, following the substitution of one ormore of the isolated bit errors for short bursts of bit errors, theblock will still be decoded error-free. The only penalty that remainsassociated with the use of differential logical detection in conjunctionwith an FEC code that inherently corrects short error bursts is thedifference between the gain of the chosen FEC code compared to the gainof the best possible code that could have been used. In practice thispenalty is very small.

The G.975 code used in fiber optic transmission is based on aReed-Solomon code, but G.975 does not lead to the absence of a penaltywhen used with differential logical detection. The reason is that G.975includes an interleaving stage before the Reed-Solomon decoder. Theinterleaving stage effectively demultiplexes the FEC block into a numberof columns of length J_(FEC), and then processes the rows for FECdecoding. The purpose of the interleaving stage is to spread out a longburst of errors, longer than the length of burst that the Reed-Solomoncode corrects naturally, so that the long error burst can be correctedwithout causing bit errors to remain after FEC decoding. An additionaladvantage of the interleaver is that the FEC block is easier to processby a parallel architecture digital processor in its demultiplexed form.The interleaving stage has the disadvantage that a short burst of errorsis also spread out, and so it does not allow short error bursts to becorrected as if they were single errors. To avoid this behavior, theinterleaver may be omitted. Alternatively the FEC code may be composedof a low overhead code that corrects short error bursts, followed by aninterleaver, and then followed by a code to correct the remainingerrors. This way the short error bursts arising from differentiallogical detection are corrected without penalty. In addition long burstsof errors are corrected, and the overall FEC code may be engineered tohave high coding gain. A second alternative is for the differentiallogical detection operation to compare with an earlier symbol instead ofthe immediately preceding symbol, and so to have output {circumflex over(d)}(n−J_(diff))*{circumflex over (d)}(n). J_(diff) is chosen to be amultiple of J_(FEC). This means that the burst of errors resulting froma single symbol error going into the differential logical detectionoperation is spread out at the output of the differential logicaldetection operation, and then appears as a short burst in one row of theinterleaved FEC block. When a cycle slip occurs it causes J_(diff) shortbursts of bit errors instead of a single short burst of bit errors, butthese short bursts appear in different FEC block rows and so arecorrected without causing excess errors.

While the present invention is described herein with reference toillustrative embodiments for particular applications, it should beunderstood that the invention is not limited thereto. Those havingordinary skill in the art and access to the teachings provided hereinwill recognize additional modifications, applications, and embodimentswithin the scope thereof and additional fields in which the presentinvention would be of significant utility.

Thus, the present invention has been described herein with reference toa particular embodiment for a particular application. Those havingordinary skill in the art and access to the present teachings willrecognize additional modifications, applications and embodiments withinthe scope thereof.

It is therefore intended by the appended claims to cover any and allsuch applications, modifications and embodiments within the scope of thepresent invention.

Appendix A. Use of Complex Numbers to Describe Modulated Signals

Complex numbers are used to describe sine and cosine functions becausethis notation is a compact way of including the phase of the sine waveor cosine wave. For example the electric field is written in the formE(t)=Re└E _(s) e ^(iωt)┘  (A1)where E_(s) is a complex number. This may be expressed in terms of sinesand cosines asE(t)=Re[E _(s)]cos(ωt)−Im[E _(s)]sin(ωt)Or if complex E_(s) is written in terms of its magnitude and phaseE _(s) =|E _(s) |e ^(iθ) ^(s)then A1 becomesE(t)=|E _(s)|cos(ωt+θ _(s))The complex number notation is compact because the phase of the sinewave is stored in the phase of the complex number.

In the above discussion are equations similar tobeat term=Re└E _(s)E_(LO) *e ^(iωt┘)  (A2)E _(LO)* is the complex conjugate of E_(LO), meaning that everyoccurrence of i is replaced with −i, andE _(LO) *=|E _(LO) |e ^(−iθ) ^(LO)A2 may be rewritten asbeat term=|E_(s) ||E _(LO)|cos(ωt+θ _(s)−θ_(LO))The appearance of E_(s)E_(LO)* in A2 means to take the phase differencebetween E_(s) and E_(LO).

The power of an optical wave is given by the magnitude squared of thecomplex electric field, and does not have a sinusoid time dependence. Soin the case of a field given by A1power=(E _(s) e ^(iωt))*(E _(s) e ^(iωt))=|E _(s)|²Appendix B. Jones Vectors

The state of polarization of an optical signal may be described by aJones vector. This is a two element column vector. Each element is thecomplex envelope of the electric field, i.e., phase informationincluded. The top element is the component of the field in thex-direction (horizontal) and the bottom element in the y-direction(vertical). In fact x and y may be an arbitrary pair of orthogonaldirections. “Optics” by E. Hecht (Addison-Wesley, 4th ed., 2001) gives athorough account of Jones vectors.

Some Jones vectors of familiar states of polarization are listed below.$\begin{matrix}\begin{pmatrix}1 \\0\end{pmatrix} & {horizontal} \\\begin{pmatrix}0 \\1\end{pmatrix} & {vertical} \\{\frac{1}{\sqrt{2}}\begin{pmatrix}1 \\1\end{pmatrix}} & {{linearly}\quad{polarized}\quad{at}\quad 45{^\circ}} \\{\frac{1}{\sqrt{2}}\begin{pmatrix}1 \\i\end{pmatrix}} & {circular}\end{matrix}$

A Jones unit vector {tilde over (p)} has the property that{tilde over (p)}·{tilde over (p)}*=1If light polarized in SOP {tilde over (p)}₁ passes through a polarizeroriented in direction {tilde over (p)}₂, then the magnitude of theelectric field is scaled by {tilde over (p)}₁·{tilde over (p)}₂*, andthe direction of the electric field is changed to {tilde over (p)}₂. Ingeneral 0≦|{tilde over (p)}₁*{tilde over (p)}₂*|≦1.

When polarized light is passed through a linear optical element, thetransformation of the SOP is described by premultiplying by a 2×2 matrixcalled the Jones matrix of the optical element.

1. A coherent optical detection system receiving an input optical signalcontaining digital information, the input optical signal having signalphase noise and additive noise, the coherent optical detection systemcomprising: a local oscillator laser, the local oscillator later havinglocal oscillator phase noise; and a digital signal processor, thedigital signal processor having a parallel architecture; wherein: thedigital signal processor applies a nonlinear function to values of thecomplex electric field of the input optical signal to produce nonlinearfunction output values; and the digital signal processor applies afilter to values derived from the nonlinear function output values, saidfilter being close to a Wiener filter appropriate to the additive noise,signal phase noise and local oscillator phase noise; whereby saiddigital signal processor estimates the phase of the input optical signalcompared to light from the local oscillator.
 2. The coherent opticaldetection system of claim 1 wherein the filter is close to a finite-lagWiener filter.
 3. The coherent optical detection system of claim 1wherein the filter is close to a zero-lag Wiener filter.
 4. The coherentoptical detection system of claim 1 wherein the Wiener filter isappropriate to Lorentzian lineshape signal source and local oscillatorlaser and appropriate to Gaussian additive noise.
 5. The coherentoptical detection system of claim 1 wherein the filter has an impulseresponse, and the non-zero values of the impulse response have a featurethat the trend of the values of the impulse response is to decay fromzero time in the direction of negative time, and to decay from zero timein the direction of positive time.
 6. A coherent optical detectionsystem receiving an input optical signal containing digital information,the input optical signal having signal phase noise and additive noise,the coherent optical detection system comprising: a local oscillatorlaser, the local oscillator laser having local oscillator phase noise;and a digital signal processor, the digital signal processor having aparallel architecture; wherein: the digital signal processor applies anonlinear function to values of the complex electric field of the inputoptical signal to produce nonlinear function output values; and thedigital signal processor applies a cycle count function to valuesderived from the nonlinear function output values; whereby said digitalsignal processor estimates the phase of the input optical signalcompared to light from the local oscillator.
 7. The coherent opticaldetection system of claim 6 wherein: said digital signal processorapplies a filter to values derived from the nonlinear function outputvalues to produce filtered output values, said filter being close to aWiener filter appropriate to the additive noise, signal phase noise andlocal oscillator phase noise; and the digital signal processor appliessaid cycle count function to values derived from said filtered outputvalues.
 8. The coherent optical detection system of claim 6 wherein:said digital signal processor calculates the phase angle of thenonlinear function output values to produce wrapped phase angle values;and the digital signal processor applies the cycle count function to thewrapped phase angle values to produce unwrapped phase angle values; andthe digital signal processor applies a filter to values derived from theunwrapped phase angle values, said filter being close to a Wiener filterappropriate to the additive noise, signal phase noise and localoscillator phase noise.
 9. A coherent optical detection system receivingan input optical signal containing digital information, the inputoptical signal having signal phase noise and additive noise, thecoherent optical system comprising: a local oscillator laser, the localoscillator laser having local oscillator phase noise; and a digitalsignal processor, the digital signal processor having a parallelarchitecture; wherein: the digital signal processor makes an estimate ofthe phase of the input optical signal compared to light from the localoscillator; and the digital signal processor includes a differentiallogical detection operation; and the data values produced by thedifferential logical detection operation are communicated to a forwarderror correction decoder; and said forward error correction decoderdecodes a forward error correction code, the forward error correctioncode being of the type that inherently corrects short bursts of errors;whereby cycle slip events caused by an imperfect estimate of the phasedo not cause a large number of bit errors to be produced by the coherentoptical detection system.
 10. The coherent optical detection system ofclaim 9 wherein: the forward error correction decoder decodes a firstforward error correction code, the first forward error correction codeinherently correcting short bursts of errors; the forward errorcorrection decoder performs an interleaving operation; the forward errorcorrection decoder decodes a second forward error correction code. 11.The coherent optical detection system of claim 9 wherein: thedifferential logical decoder compares each symbol with another symbolwhich is not an immediately preceding symbol; and the forward errorcorrection decoder performs an interleaving operation; and the forwarderror correction decoder decodes the forward error correction code, theforward error correction code inherently correcting short bursts oferrors.
 12. A method of estimating the phase of an input optical signalto a coherent optical receiver compared to light from a local oscillatorlaser, said method comprising the steps of: applying a nonlinearfunction to values of the complex electric field of the input opticalsignal to produce nonlinear function output values; and applying adigital filter to values derived from the nonlinear function outputvalues, said digital filter being close to a Wiener filter appropriateto the additive noise and the phase noise on the input optical signaland the phase noise of the local oscillator, and said digital filter notemploying feedback of immediately preceding results.
 13. The method ofclaim 12 wherein the digital filter is close to a zero-lag Wiener filterappropriate to Lorentzian lineshape signal source and Lorentzianlineshape local oscillator laser and appropriate to Gaussian additivenoise.
 14. The method of claim 12 wherein the digital filter is close toa finite-lag Wiener filter appropriate to Lorentzian lineshape signalsource and Lorentzian lineshape local oscillator laser and appropriateto Gaussian additive noise.
 15. A method of estimating the phase of aninput optical signal to a coherent optical receiver compared to lightfrom a local oscillator laser, said method comprising the steps of:applying a nonlinear function to values of the complex electric field ofthe input optical signal to produce nonlinear function output values;and applying a cycle count function to values derived from the nonlinearfunction output values, the cycle count function not employing feedbackof immediately preceding results.
 16. The method of claim 15 furthercomprising the steps of: applying a digital filter to values derivedfrom the nonlinear function output values to produce digital filteroutput values, said digital filter being close to a Wiener filterappropriate to the additive noise and the phase noise on the inputoptical signal and the phase noise of the local oscillator, said digitalfilter not employing feedback of immediately preceding results; andapplying the cycle count function to values derived from the digitalfilter output values.
 17. A method of obtaining information carried onan input optical signal received by an optical coherent detectionsystem, the method comprising the steps of: estimating the phase of theinput optical signal compared to a local oscillator; and applying thephase estimate to produce symbol output values; applying a differentiallogical detection operation to the symbol output values to producedifferential symbol values; and decoding a forward error correction codeon the differential symbol values, the forward error correction codeinherently correcting short bursts of errors.
 18. The method of claim 17wherein the following steps are performed subsequent to the step ofdecoding the forward error correction code that inherently correctsshort bursts of errors: performing an interleaving operation; anddecoding a second forward error correction code.
 19. The method of claim17 further comprising the following step of: performing an interleavingoperation on the differential symbol values; and wherein thedifferential logical detection includes comparing a symbol output valuewith another symbol output value which is not the immediately precedingsymbol output value.